Ultra-wideband communication systems and methods

ABSTRACT

Systems and methods of ultra-wideband communication are provided. In one embodiment, an ultra-wideband communication system divides a stream of data conveying symbols into a plurality of unspread substreams. A common spreading code is generated at the ultra-wideband transmitter, and each of the unspread substreams are spread using the common spreading code to form a plurality of spread substreams. The spread substreams are combined to form a composite signal that is transmitted. This Abstract is provided for the sole purpose of complying with the Abstract requirement rules that allow a reader to quickly ascertain the subject matter of the disclosure contained herein. This Abstract is submitted with the explicit understanding that it will not be used to interpret or to limit the scope or the meaning of the claims.

This application claims priority under 35 U.S.C. § 120 as a continuationof co-pending U.S. application Ser. No. 10/961,592, filed Oct. 8, 2004,entitled “ULTRA-WIDEBAND COMMUNICATION SYSTEMS AND METHODS,” which is acontinuation-in-part of U.S. application Ser. No. 09/670,054 filed Sep.25, 2000 entitled “METHOD AND APPARATUS FOR WIRELESS COMMUNICATIONS,”now U.S. Pat. No. 7,031,371.

TECHNICAL FIELD OF THE INVENTION

The invention relates generally to ultra-wideband communications, andmore particularly to systems and methods for communication usingultra-wideband technology.

BACKGROUND OF THE INVENTION

The electromagnetic spectrum used to convey radio communications is aprecious commodity. Communication systems seek to use this spectrum asefficiently as possible to maximize the capacity or quantity ofinformation, which can be conveyed using the spectrum.

Various multiple access techniques have been developed to transferinformation among a number of users, all while efficiently usingspectrum. Time division multiple access (TDMA) techniques assigndifferent users to different time slots. Capacity is hard limited by thenumber of time slots available. To prevent intolerable interference, theportion of the spectrum used in one radio coverage area or cell hasconventionally been unusable in adjacent cells. Thus, only a fraction,typically less than one-third, of the entire spectrum available forconveying communications has been conventionally usable in any onelocation. In other words, conventional TDMA systems employ a frequencyreuse pattern of at least three, indicating an inefficient use ofspectrum.

Conventional direct sequence spread spectrum (DSSS) code divisionmultiple access (CDMA) techniques theoretically use the spectrum moreefficiently than TDMA techniques. However, in practice conventionalDSSS-CDMA techniques typically fail to provide results significantlybetter than TDMA. DSSS-CDMA techniques assign different users todifferent codes. The different codes have conventionally been selectedbecause of orthogonality or low cross correlation properties with thecodes of other users. These properties minimize interference. Allcommunications are broadcast using the same spectrum, so the frequencyreuse pattern equals one. While the commonly used spectrum conveys acomposite of communications for all users, each individual user'scommunications are extracted from the composite by correlating areceived signal against the individual user's assigned code.

Capacity in conventional DSSS-CDMA systems is interference limited. Inother words, more and more codes can be assigned so that the givenamount of spectrum can service more and more users until interferencereaches a level where only a minimally acceptable quality of serviceresults. In practice, most conventional DSSS-CDMA systems can assign farfewer codes than appear theoretically possible due to a near-far effectand multipath. The near-far effect results when signals from differentusers are received with greatly differing field strengths, but thisdetrimental effect may be ameliorated somewhat by power control.

Multipath results when the transmitted signal takes multiple paths tothe receiver due to being reflected from and deflected around obstaclesin the environment. As the signal propagates over the multiple paths,different propagation delays are experienced. Thus, a signal transmittedat a precise instant in time is received spread over an interval,causing the signal to interfere with itself. In conventional DSSS-CDMAcommunication systems, multipath tends to destroy the orthogonality ofspreading codes, resulting in dramatically increased interference.

SUMMARY OF THE INVENTION

In order to combat the above problems, systems and methods describedherein provide a novel ultra-wideband communication system. In oneembodiment, an ultra-wideband communication system divides a stream ofdata conveying symbols into a plurality of unspread substreams. A commonspreading code is generated at the ultra-wideband transmitter, and eachof the unspread substreams are spread using the common spreading code toform a plurality of spread substreams. The spread substreams arecombined to form a composite signal that is transmitted.

In another embodiment, an ultra-wideband communication system comprisesa demultiplexer for dividing a stream of data conveying symbols into aplurality of unspread substreams. A spreading section is coupled to thedemultiplexer and configured to generate a plurality of spreadsubstreams from the plurality of unspread substreams. A combiningsection is coupled to the spreading section and configured to form acomposite signal from the plurality of spread substreams, and atransmission section is coupled to the combining section and configuredto transmit the composite signal over an ultra-wideband communicationchannel.

These and other features and advantages of the present invention will beappreciated from review of the following Detailed Description of thePreferred Embodiments, along with the accompanying figures in which likereference numerals are used to describe the same, similar orcorresponding parts in the several views of the drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

A more complete understanding of the present invention may be derived byreferring to the detailed description and claims when considered inconnection with the Figures, wherein like reference numbers refer tosimilar items throughout the Figures, and:

FIG. 1 shows a layout diagram of an exemplary environment in which thepresent invention may be practiced;

FIG. 2 shows a timing diagram, which depicts a temporal format of a TDMAcommunication signal;

FIG. 3 shows a block diagram of a transmitter and a receiver configuredin accordance with the teaching of the present invention;

FIG. 4 shows a timing diagram depicting how a cyclic spreading code isapplied to blocks of unspread data streams in accordance with first,second and third embodiments of a DSSS modulation section in thetransmitter of the present invention;

FIG. 5 shows a block diagram of the first embodiment of the DSSSmodulation section;

FIG. 6 shows a block diagram of the second embodiment of the DSSSmodulation section;

FIG. 7 shows a block diagram of the third embodiment of the DSSSmodulation section;

FIG. 8 shows a first embodiment of a CDM to TDM converter section in thereceiver of the present invention;

FIG. 9 shows an exemplary spectral analysis of a suitable spreading codeusable in connection with the present invention, the spectral analysisshowing a substantially flat response;

FIG. 10 shows an exemplary timing diagram of various individual signalcomponents present in a composite signal output from a matched filterportion of a mismatched filter in the CDM to TDM converter;

FIG. 11 shows a timing diagram depicting how a cyclic spreading code isapplied to blocks of unspread data streams in fourth and fifthembodiments of the DSSS modulation section;

FIG. 12 shows a block diagram of the fourth and fifth embodiments of theDSSS modulation section;

FIG. 13 shows a second embodiment of the CDM to TDM converter for usewith the fourth embodiment of the DSSS modulation section;

FIG. 14 shows a third embodiment of the CDM to TDM converter for usewith the fifth embodiment of the DSSS modulation section;

FIG. 15 is an illustration of different communication methods; and

FIG. 16 is an illustration of two ultra-wideband pulses.

It will be recognized that some or all of the Figures are schematicrepresentations for purposes of illustration and do not necessarilydepict the actual relative sizes or locations of the elements shown. TheFigures are provided for the purpose of illustrating one or moreembodiments of the invention with the explicit understanding that theywill not be used to limit the scope or the meaning of the claims.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

In the following paragraphs, the present invention will be described indetail by way of example with reference to the attached drawings. Whilethis invention is capable of embodiment in many different forms, thereis shown in the drawings and will herein be described in detail specificembodiments, with the understanding that the present disclosure is to beconsidered as an example of the principles of the invention and notintended to limit the invention to the specific embodiments shown anddescribed. That is, throughout this description, the embodiments andexamples shown should be considered as exemplars, rather than aslimitations on the present invention. As used herein, the “presentinvention” refers to any one of the embodiments of the inventiondescribed herein, and any equivalents. Furthermore, reference to variousfeature(s) of the “present invention” throughout this document does notmean that all claimed embodiments or methods must include the referencedfeature(s).

The present invention provides several advantages and features, forexample, the present invention combines TDMA and spread spectrumtechniques so that wireless communications capacity is increased overthe capacities achievable through conventional TDMA and/or CDMA systemsusing an equivalent amount of spectrum.

Another advantage of the present invention is that robust, simple, andinexpensive processing techniques are usable, making the presentinvention suitable for hubs, subscriber units, mobile stations/fixedstations, portable stations, and the like.

Another advantage is that the present invention may be adapted to andused in conjunction with a variety of modulation and multiple accesstechniques, such as frequency division multiple access (FDMA) andorthogonal frequency division multiplexing (OFDM).

Another advantage of the present invention is that a composite RFcommunication signal includes signal components obtained by modulatingdiverse branches of a single user's data stream using cyclic variants ofa common spreading code.

Another advantage is that the present invention is configured totolerate self-interference and is better able to tolerate multipath thanconventional DSSS-CDMA communication systems.

These and other features and advantages of the present invention will beappreciated from review of the following discussion:

FIG. 1 shows a layout diagram of an exemplary environment in which acommunication system 20 configured in accordance with the teaching ofthe present invention may be practiced. Communication system 20 includesany number of transmitters (TX's) 22 (three shown) and any number ofreceivers (RX's) 24 (five shown). Transmitters 22 wirelessly broadcastmessages through RF time domain multiple access (TDMA) communicationsignals 26 which are receivable by receivers 24 located within radiocoverage areas 28 for the transmitters 22. Radio coverage areas 28 mayalso be called cells or sectors. As illustrated in FIG. 1, various onesof radio coverage areas 2 8 may be adjacent to one another and evenoverlap to some extent. In the preferred embodiment, a common spectrumis used in all radio coverage areas 28 so that communication system 20has a frequency reuse pattern substantially equal to one.

For the sake of clarity, FIG. 1 depicts only a forward link in whichradio equipment is viewed as being only a transmitter 22 or a receiver24. However, those skilled in the art will appreciate that a reverselink may also be implemented and that each item of equipment may haveboth a transmitter and receiver. The reverse link may use the same or adifferent spectrum from the forward link. If a forward link conforms tothe teaching of the present invention, then the reverse link may or maynot conform, and vice versa.

FIG. 2 shows a timing diagram, which depicts an exemplary temporalformat for TDMA communication signal 26. FIG. 2 specifically depicts twoframes 30, each of which is temporally subdivided into any number oftimeslots 32. Different timeslots 32 are preferably assigned todifferent receivers 24 (FIG. 1) in a manner well understood in the artso that different recipients are distinguished from one another by beingassigned to the different time slots 32. In the preferred embodiments,TDMA communication signal 26 consumes the entire common spectrum foreach time slot 32. Nothing requires a time slot 32 to be assigned toreceivers 24 for an indefinite period or to be of the same duration asother time slots 32.

Each time slot 32 of TDMA communication signal 26 is subdivided intosuccessive blocks 34 of symbols 36. FIG. 2 labels blocks 34 with theidentifiers B_(k), for k=0 to K−1, where K is an integer number. Anynumber of blocks 34 may be included in each timeslot 32. Each blockB_(k) includes M symbols 36, labeled as a_(k,m) for m=0 to M−1, where Mis an integer number. FIG. 2 illustrates each of symbols 36 within ablock 34 as being concurrently present throughout the entire duration ofa block period because certain preferred embodiments discussed belowconfigure symbols 36 to remain present for block periods.

FIG. 2 further illustrates that the M symbols 36 of each block 34 arespread using an N-chip spreading code 38, labeled as C_(n), for n=0 toN−1, where N is an integer number. As discussed in more detail below,each symbol 36 is independently spread using cyclic variations of thesame common code 38. The number M of symbols 36 in a block may equal thenumber N of chips in a spreading code, in which case the spreadingfactor equals one. However, performance improvements result when N isgreater than M.

FIG. 3 shows a block diagram of a single transmitter 22 and a singlereceiver 24 configured in accordance with the teaching of the presentinvention. Those skilled in the art will appreciate that alltransmitters 22 and receivers 24 may be configured similarly. Inaddition, any number of receivers 24 may, at any given instant, receiveTDMA communication signal 26 from a given transmitter 22 and, in fact,may receive TDMA communication signals 26 from more than one transmitter22.

Transmitter 22 includes a TDMA modulation section 40, which generates aTDMA-configured stream 42 of data conveying symbols 36. Stream 42 feedsa direct sequence spread spectrum (DSSS) modulation section 44, whichgenerates a composite signal 46. Composite signal 46 feeds atransmission section 48, which forms TDMA communication signal 26 fromcomposite signal 46 and wirelessly broadcasts TDMA communication signal2 6 for reception by receivers 24 located within radio coverage area 28(FIG. 1) of transmitter 22

Within TDMA modulation section 40 any number of data sources 50 supplydigital data to a multiplexer (MUX) 52. The digital data from datasources 50 may be intended for any number of receivers 24. Multiplexer52 groups the digital data so that data intended for different receivers24 are serially fed to a cyclic redundancy check (CRC) section 54 inaccordance with the assignment of timeslots 32 (FIG. 2) to receivers 24.CRC section 54 provides forward error correction in a manner wellunderstood by those skilled in the art.

From CRC section 54, the input data stream may be fed through ascrambler 56 which randomizes the data to an encode and interleavesection 58. Section 58 may apply another type of error correction, suchas convolutional or turbo encoding, to the input stream, and interleavethe data. CRC section 54 and section 58 may utilize a form of blockencoding. The block size or boundaries of such encoding need have norelationship to blocks 34 (FIG. 1), discussed above.

However, the output of section 58 feeds an optional peak-to-average(P/A) block encoding section 60. P/A block encoding section 60 applies atype of encoding which primarily reduces the peak-to-average power ratioin composite signal 46 and thereby lessens the demands placed on a poweramplifier included in transmission section 48 to faithfully reproducecommunication signal 26 with a minimum amount of distortion. This typeof encoding may, but is not required to, provide additional coding gain.In the preferred embodiments, when P/A block encoding section 60 isincluded, it applies block encoding so that encoded blocks coincide withsuccessive blocks of symbols 36 (FIG. 2), discussed above. In otherwords, the data are encoded so that P/A encoded blocks begin with symbol36 a_(k,0) (FIG. 2) and end with symbol 36 a_(k,m) (FIG. 2).

P/A block encoding section 60 feeds a constellation encoding section 62which converts the data into complex symbols in accordance with apredetermined phase constellation. As an example, each four-bit group ofdata output from P/A block encoding section 60 may be mapped by section62 into a single complex symbol having in-phase and quadraturecomponents in accordance with a 16-QAM phase constellation. However,those skilled in the art will appreciate that the present invention maybe used with any type or size of phase constellation.

The stream of complex symbols output from constellation encode section62 passes through a synchronization multiplexer (SYNC MUX) 64, where apreamble 66 is inserted into the stream at appropriate intervals.Preamble 66 is a known code which helps receivers 24 obtainsynchronization and determine the timing of frames 30 and time slots 32(FIG. 2). The resulting TDMA-configured complex stream 42 serves as theoutput from TDMA modulation section 4 0 and feeds DSSS modulationsection 44.

Within DSSS modulation section 44, a demultiplexer (DEMUX) 68 dividesTDMA-configured stream 42 of complex symbols 36 into blocks 34 (FIG. 2)of symbols 36. As a result, M unspread complex symbol substreams 70 areprovided by demultiplexer 68 so that each unspread substream 70contributes a single complex symbol 3 6 during each block 34, and eachblock 34 has a block period T*M, where T is the symbol period ofTDMA-configured stream 42.

Unspread substreams 70 feed a spreading section 72. Within spreadingsection 72, cyclic variations of common spreading code 38 (FIG. 2) areapplied to the M unspread substreams 7 0 to form M spread substreams 74of “chips.” The chip period in each spread substream 74 is T*M/N. The Mspread substreams 74 may be passed through an optional peak-to-average(P/A) reduction section 76 which adjusts phase angles of the complexchips conveyed in the spread substreams 74 in a manner understood bythose skilled in the art to reduce peak-to-average power ratio andlessen demands placed on a power amplifier. Following P/A reductionsection 76, a combining section 78 combines spread substreams 74 to formcomposite signal 46. Various embodiments of DSSS modulation section 44are discussed in more detail below.

Transmission section 48 includes any number of components and functionswell known to those skilled in the art. For example, scrambling section56 and/or synchronization multiplexer 64, discussed above, may beincluded in transmission section 48 rather than in TDMA modulationsection 40. A pulse shaping section (not shown) is desirably included intransmission section 48 to spread the energy from each chip over anumber of chip intervals using a suitable filter which minimizesinter-symbol or inter-chip interference so that spectral constraints maybe observed. Transmission section 48 may also include digital-to-analogconversion, quadrature modulation, up-conversion, and poweramplification functions, all implemented in conventional fashion. Powercontrol may be implemented in transmission section 48 at the poweramplifier to ameliorate a potential near-far problem, which should bemuch less pronounced in communication system 2 0 (FIG. 1) than intraditional CDMA communication systems. After pulse shaping, analogconversion, up-conversion, and amplification, TDMA communication signal2 6 is formed from composite signal 4 6 and wirelessly broadcast fromtransmission section 48. Receiver 24 receives TDMA communication signal26.

Within receiver 24, communication signal 26 is processed through areceiving section 80 and passed to a code division multiplex (CDM) totime division multiplex (TDM) converter 82. CDM to TDM converter 82produces a baseband signal 84, which is further demodulated in a TDMdemodulation section 86, with individual users receiving theirrespective data streams 88. Of course, nothing requires a receiver 24 toserve multiple users and TDM demodulation section 86 may simply providea data stream intended for a single user.

Receiving section 80 includes any number of components and functionswell known to those skilled in the art. For example, amplifying,filtering, and down-conversion may be performed to form an intermediatefrequency (IF) signal. The IF signal may be converted from an analogform into a digital form, and automatic gain control (AGC) may beprovided. In the preferred embodiments, the digitized form of thedown-converted communication signal 2 6 passes to CDM to TDM converter82.

Generally, CDM to TDM converter 82 performs despreading and optionallyperforms equalization on the communication signal. Various embodimentsof CDM to TDM converter 82 are discussed in more detail below.

TDM demodulation section 86 includes any number of components andfunctions well known to those skilled in the art. For example, channelestimation and synchronization may be performed in TDM demodulationsection 82. A rake receiver and/or equalizer may be included.De-interleaving, error correction decoding, and descrambling aredesirably performed, and preambles and other control data are evaluatedto detect time slots assigned to the receiver 24. These and othercomponents and functions conventionally used in digital demodulators maybe included in TDM demodulation section 86.

FIG. 4 shows a timing diagram depicting how common spreading code 38(FIG. 2) is applied to blocks 34 (FIG. 2) of unspread data substreams 70(FIG. 3) in accordance with first, second and third embodiments of DSSSmodulation section 44 (FIG. 3) in transmitter 22 (FIG. 3). FIG. 4 ispresented in tabular form, with rows representing the application of thechips of common spreading code 38 to symbols 36 (FIG. 2). Columns inFIG. 4 depict successive blocks 34. As indicated by a shaded region inFIG. 4, spreading code 38 is applied to unspread substreams 70 so thatcomposite signal 46 is influenced, at least for a portion of the time,by symbols 36 from two different blocks 34. In particular, in thespecific embodiment depicted by FIG. 4, for only a single chip of eachblock period is composite signal 4 6 influenced by symbols 3 6 from acommon block 34 of symbols. The manner of application of commonspreading code 38 (FIG. 2) to blocks 34 (FIG. 2) of unspread datasubstreams 70 depicted in FIG. 4 may be contrasted with an alternateembodiment, discussed below in connection with FIG. 11.

FIG. 5 shows a block diagram of the first embodiment of DSSS modulationsection 44. Demultiplexer 68 is omitted from FIG. 5 for convenience. Theunspread substream 70 conveying symbols a_(k,0) experiences no delaybefore being fed to a first input of a multiplier 90. However, theunspread substreams 70 conveying symbols a_(k,1) through a_(k,M-1) arerespectively delayed in delay elements 92 by 1 through M−1 symbolperiods (T) before being fed to respective first inputs of othermultipliers 90.

A spreading code generation section 94 generates cyclic variations ofcommon spreading code 38. FIG. 5 illustrates code generation section 94in matrix form, which matrix takes on a cyclic Toeplitz form because thematrix elements hold cyclic variations of the same spreading code 38. Asdepicted in FIG. 5, different columns of the matrix supply code chips COthrough CN−1 to second inputs of respective multipliers 90. Differentrows of the matrix indicate different code chips to apply duringdifferent chip intervals. So long as the number (N) of chips inspreading code 38 is greater than or equal to the number (M) of symbols36 per block 34, different code chips of the same code are applied todifferent symbols during any and all chip intervals.

Outputs of multipliers 90 provide respective spread substreams 74. FIG.5 omits depiction of optional P/A reduction section 76 (FIG. 3) forconvenience. Combining section 78 takes the form of an adder, so thatcomposite signal 4 6 during each chip interval equals the sum of Msymbols 36, with each of the M symbols being premultiplied by designatedchips of common spreading code 38. Accordingly, DSSS modulating section44 temporally offsets application of common spreading code 38 tounspread substreams 70 so that the resulting spread substreams 74correspond to unspread substreams 70 modulated by cyclic variations ofcommon spreading code 38.

FIG. 6 shows a block diagram of the second embodiment of DSSS modulationsection 44. This second embodiment is equivalent to the first embodimentof FIG. 5, but it is implemented differently. Demultiplexer 68 (FIG. 3)is omitted from FIG. 6 for convenience. In this embodiment, spreadingcode generation section 94 need not be implemented as a two-dimensionalmatrix having a different row to define the different chips to beapplied during different chip intervals, as discussed above inconnection with FIG. 5. Rather, spreading code generation section 94 maybe implemented as a one-dimensional matrix having different columns, andonly one of those columns is simultaneously applied to differentunspread substreams 70. Spreading code generation section 94 may beimplemented as a shift register configured to shift cyclically at thechip rate. In order to achieve the appropriate temporal offsetting,delay elements 92 are now positioned between multipliers 90 and theadder of combining section 78. Accordingly, in this second embodiment ofDSSS modulating section 44, DSSS modulating section 44 temporallyoffsets the application of common spreading code 38 to unspreadsubstreams 70 so that the resulting spread substreams 74 correspond tounspread substreams 70 modulated by cyclic variations of commonspreading code 38.

FIG. 7 shows a block diagram of the third embodiment of DSSS modulatingsection 44. This third embodiment is also equivalent to the firstembodiment of FIG. 5, but is implemented differently. This thirdembodiment is a finite impulse response (FIR) implementation. In thisthird embodiment, symbol stream 42 (FIG. 3) is fed to a series of delayelements 96, each of which imparts a one-chip interval delay. The seriesof delay elements 96 serves the role of demultiplexer 68 (FIG. 3) inthis third embodiment, with the input to the first delay element 96 andthe outputs of all delay elements 96 providing unspread substreams 70.Delay elements 92 (FIGS. 5-6) from the first and second embodiments ofspreading section 72 are omitted.

Spreading code generation section 94 simply provides common spreadingcode 38, and need not be cycled because unspread substreams 70 to whichspreading code 38 is applied are configured to perform the temporaloffsetting requirements. Accordingly, symbol delay elements 92 areomitted and spreading code generating section 94 need not cycle thecommon spreading code or explicitly provide separate versions ofspreading code 3 8 to separate unspread substreams 70. Nevertheless, inthis third embodiment of DSSS modulation section 44, spreading section72 temporally offsets application of common spreading code 3 8 bysequentially delaying symbols 36 to form unspread substreams 70 andapplying spreading code 38 to the delayed symbols in unspread substreams70 so that the resulting spread substreams 74 correspond to unspreadsubstreams 70 modulated by cyclic variations of common spreading code38.

FIG. 8 shows a first embodiment of CDM to TDM converter 82 included inreceiver 24 (FIGS. 1 and 3). Desirably, CDM to TDM converter 82 isconfigured to complement DSSS modulation section 44 of transmitter 22(FIGS. 1 and 3). In particular, this first embodiment of CDM to TDMconverter 82 is configured to complement any of the first through thirdembodiments of DSSS modulation section 44 discussed above in connectionwith FIGS. 4-7.

CDM to TDM converter 82 includes a pulse shaping matched filter 98, theoutput of which feeds a mismatched filter 100. Pulse shaping matchedfilter 98 complements a pulse shaping filter (not shown) desirablyimplemented in transmission section 48 of transmitter 22 (FIG. 3) tooptimize signal-to-noise ratio and band-limit the signal. Pulse shapingmatched filter 98 is desirably implemented using conventional techniquesknown to those skilled in the art.

Mismatched filter 100 accomplishes two functions. One function isdespreading and the other function is sidelobe suppression. In fact,mismatched filter 100 is desirably implemented to correspond to aspreader matched filter 102 upstream of a sidelobe suppression filter104. One technique for implementing mismatched filter 100 is simply toimplement two filters coupled in series for the despreading and sidelobesuppression functions. In another technique (not shown) the twofunctions may be combined in a common filter.

Mismatched filter 100 experiences a signal-to-noise ratio typicallyworse than that of a matched filter. However, in the preferredembodiments, mismatched filter 100 is desirably configured to achieve arelative efficiency of greater than 60%, and more preferably greaterthan 90%, compared to a matched filter.

Those skilled in the art will appreciate that the configuration ofcommon spreading code 3 8 is a strong determinant of the relativeefficiency of mismatched filter 100. For example, conventionalorthogonal pseudonoise (PN) codes commonly used in conventional CDMAapplications are unacceptable because their mismatched filters achieverelative efficiencies roughly around only 50%.

While a wide variety of different codes may be used with the presentinvention, codes which have low aperiodic autocorrelation sidelobes anda substantially flat spectral analysis are preferred in this embodiment.Barker codes make suitable codes because of aperiodic autocorrelationsidelobes having magnitudes less than or equal to one. However, for manyapplications the limited length (i.e., N≦13) and/or prime numberedlength of many Barker codes proves a detriment. In such cases, othercodes having a greater length and slightly greater aperiodicautocorrelation sidelobes, such as magnitudes less than or equal to twoor three are acceptable and may be easily derived by those skilled inthe art.

FIG. 9 shows an exemplary spectral analysis of a suitable spreading codeusable in connection with the present invention. In particular, FIG. 9represents an arbitrary code for which a spectral analysis can beperformed using a time-frequency domain transformation, such as aFourier transform. While a code having a precisely flat spectralanalysis result is not a requirement, better results are achieved whenno frequency bin shows substantially more or less signal level thanother bins, as depicted in FIG. 9. As an example, the signal level ineach bin is desirably within ±25% of the average signal level taken overall bins. In particular, for best results no bins should exhibit anearly zero signal level.

The implementation of mismatched filter 100 illustrated in FIG. 8 willbe readily understood by those skilled in the art. Spreader matchedfilter 102 may be implemented using the complex conjugate of spreadingcode 38 (FIG. 2) presented in a reverse order. Sidelobe suppressionfilter 104 may be implemented using well-known FFT or linear programmingtechniques.

The output of spreader matched filter 102 in mismatched filter 100 is acomposite signal 106 equivalent to the autocorrelation function appliedto each of the M unspread and spread substreams 70 and 74 (FIGS. 3 and5-7) discussed above.

FIG. 10 shows an exemplary timing diagram of the various individualsignal components present in composite signal 106 output from thematched filter 102 portion of mismatched filter 100. For convenience,FIG. 10 depicts an exemplary situation where the number of substreams 70and 74 (i.e., M) equals seven and the number (i.e., N) of chips inspreading code 38 equals seven. Thus, each row in FIG. 10 represents oneof the seven substreams, and each row depicts autocorrelation with anassumed rectangular pulse. Of course, composite signal 106 is the sum ofall rows in FIG. 10 rather than seven distinct signals.

Assuming ideal synchronization where samples are taken at the integralchip intervals of 0, 1, 2 . . . , then in this example, seven successivesamples yield the signal levels of the seven substreams. However, eachsample in composite signal 106 is corrupted by self-interference 108,caused by sidelobes of the autocorrelation function. Accordingly,sidelobe suppression filter 104 (FIG. 8) substantially attenuates theself-interference 108 of the sidelobes while not severely attenuatingthe autocorrelation peak.

Referring back to FIGS. 3 and 8, the output sidelobe suppression filter104 provides baseband signal 84, which also serves as the output of CDMto TDM converter 82. Baseband signal 84 is routed to TDM demodulationsection 86. Depending upon the severity of multipath remaining inbaseband signal 84 after processing through sidelobe suppression filter104, a rake receiver (not shown) or equalizer (not shown) may be used inTDM demodulation section 86 to compensate for the multipath. While someinefficiency may result from using a mismatched filter to despreadcommunication signal 26, any such inefficiency is more than compensatedfor by a marked improvement in multipath tolerance.

While receiver 24 receives a communication signal 26 from onetransmitter 22, it may simultaneously receive other communicationsignals 26 from other transmitters 22 in adjacent radio coverage areas 28 (FIG. 1). Conventional CDMA techniques may be used to preventinterference between such diverse communication signals 26. For example,different spreading codes 38 may be selected for use at differenttransmitters 22. Such different spreading codes 38 are desirablyconfigured to have low cross-correlation sidelobes among all spreadingcodes 38. If this option is selected, only a few (e.g., 3-7) of suchcodes need be used to prevent interference because interference shouldnot be a problem between communication signals 26 from non-adjacentradio coverage areas 28 (FIG. 1). Alternatively, transmitter 22 andreceiver 24 may include other stages to further scramble/descramblespread spectrum signals using other spreading codes.

The embodiments of the present invention discussed above andcharacterized by the timing depicted in FIG. 4, wherein composite signal46 is influenced, at least for a portion of the time, by symbols 3 6from two different blocks 34, show advantageous resilience in thepresence of multipath. However, alternate embodiments, discussed below,may provide even better performance in the presence of multipath forsome applications.

FIG. 11 shows a timing diagram depicting how common spreading code 38(FIG. 2) is applied to blocks 34 (FIG. 2) of unspread data substreams 70(FIG. 3) in accordance with fourth and fifth embodiments of DSSSmodulation section 44 (FIG. 3) in transmitter 22 (FIG. 3). Like FIG. 4discussed above, FIG. 11 is presented in tabular form, with rowsrepresenting the application of the chips of common spreading code 38 tosymbols 36 (FIG. 2). Columns in FIG. 11 depict successive blocks 34. Asindicated by a shaded region in FIG. 11, spreading code 38 in the fourthand fifth embodiments is applied to unspread substreams 70 so thatcomposite signal 46 is influenced, at all times, by symbols 36 from onlycommon blocks 34.

FIG. 12 shows a block diagram of the fourth and fifth embodiments ofDSSS modulation section 44. Demultiplexer 68 (FIG. 3) is omitted fromFIG. 12 for convenience. In addition, in order to enable matrixmultiplication operations discussed below, FIG. 12 represents that Nsymbols 36 (i.e. a_(k,0) through a_(k,N-1)) are provided fromdemultiplexer 68 during each block 34 (FIG. 2). In other words, FIG. 12represents that the number (M) of symbols 36 per block 34 equals thenumber (N) of chips in spreading code 38. Those skilled in the art willappreciate that when M<N, the number M of symbols 3 6 per block 34 maybe made equal to N by padding with zeros so that the zeros are evenlydistributed among the symbols 36. As an example, if M equals 4 and Nequals 12, then 12 symbols 36 may be provided by following each symbol36 in each block 34 with two zeros.

Unspread substreams 70, which provide N symbols 3 6 per block 34, passto an optional time-frequency domain transformation section 110.Time-frequency domain transformation section 110 may be implemented asan inverse fast Fourier transform (IFFT). For purposes of the presentdiscussion, the fourth embodiment of DSSS modulation section 44 shall bedeemed to omit section 110, while the fifth embodiment shall be deemedto include section 110. Thus, unspread substreams 70 convey time domaindata to spreading section 72 in the fourth embodiment and frequencydomain data to spreading section 72 in the fifth embodiment.

While section 110 is not a requirement of the present invention, certainbenefits may be achieved by the addition of section 110 as will bediscussed below. Moreover, section 110, or the equivalent, isconventionally included in digital communication transmitters whichimplement an orthogonal frequency division multiplexed (OFDM) modulationformat. In such situations, section 110 may be present for use inconnection with the present invention at little additional complexity orexpense.

Delay elements 92 (FIGS. 5-6) are omitted in the fourth and fifthembodiments of DSSS modulation section 44 to permit only symbols 36concurrently present during a common block 34 to influence compositesignal 46. However, spreading section 72 and spreading code generatingsection 94 are implemented in a manner similar to that discussed abovein connection with the first and second embodiments of DSSS modulationsection 44 (FIGS. 5-6). In particular, cyclic variations of a singlecommon spreading code 38 are applied in the form of a cyclic Toeplitzmatrix (see FIG. 5). While spreading code generating section 94 acts tomultiply the 1×N matrix of symbols 36 in each block 34 by spreading code38 effectively in the form of an N×N cyclic Toeplitz matrix, it may doso simply through a one-dimensional matrix having different columnsapplied to different unspread substreams 70 at different multipliers 90.Instead of selecting a spreading code 38 with low aperiodicautocorrelation sidelobes as discussed above in connection with thefirst, second and third embodiments of DSSS modulation section 44, thefourth and fifth embodiments of DSSS modulation section 44 are betterserved with a spreading code 3 8 having low periodic autocorrelationsidelobes and a substantially flat spectrum. Spreading code generationsection 94 may be implemented as a shift register configured to shiftcyclically at the chip rate. Spread substreams 74 output frommultipliers 90 are combined in an adding circuit 78 to form apre-composite signal 46′, which is converted back into parallel streamsat a demultiplexer (DEMUX) 112 into N chips per block 34, labeledb_(k,0) through b_(k,N-1) in FIG. 12.

Demultiplexer 112 provides one technique for forming a cyclic prefix114. Chips b_(k,0) through b_(k,N-1) and cyclic prefix 114 are routed inparallel to inputs of a multiplexer (MUX) 116 for conversion into serialcomposite signal 46. In particular, chips b_(k,0) through b_(k,N-1) areassociated with an intended order, in which chips b_(k,0), b_(k,1),b_(k,2), . . . b_(k,P) occur first in pre-composite signal 46′, andchips b_(k,q), . . . b_(k,N-3), b_(k,N-2), b_(k,N-1), occur last inpre-composite signal 46′. FIG. 12 illustrates an example where the p=2first-occurring spread substreams in pre-composite signal 46′ arerepeated as cyclic prefix 114 so that they also occur last in compositesignal 46. Of course, those skilled in the art will appreciate that theclock rate of multiplexer 116 is desirably sufficiently higher than theclock chip rate to accommodate cyclic prefix 114.

Transmission section 48 forms blocks 34 of communication signal 26 fromblocks 34 of composite signal 46. Blocks 34 of communication signal 26propagate to receiver 24 through a communication channel, which may beunique to a specific transmitter 22 location and receiver 24 location.Blocks 34 of communication signal 26 experience multipath and othertypes of distortion when propagating through this channel. Themathematical effect of this distortion is equivalent to multiplyingcomposite signal 46 by the transfer function of the channel, whichimposes the multipath.

As discussed above, each block 34 of composite signal 46 is formed fromthe matrix multiplication of the spreading code 38 with a block 34 ofsymbols 36. The effect of multipath distortion is then the matrixmultiplication of the matrix expression of the channel transfer functionwith this matrix product. Normally, a matrix multiplication does notobserve a communicative mathematical property. In other words, theproduct of the channel transfer function by the spreading matrix doesnot necessarily equal the product of spreading matrix by the channeltransfer function.

Due to the failure of the mathematical communicative property in matrixmultiplication, normally equalization to compensate for multipath shouldbe performed before despreading in receiver 22. Unfortunately, suchequalization is exceedingly difficult to successfully perform, due atleast in part to requiring the implementation of a filter withcharacteristics equivalent to the inverse of the channel transferfunction. The characteristics of the channel cannot be easilycontrolled, and channel transfer function quite possibly has elementsnear zero. Attempting to form inverse filters of such characteristicsoften leads to unstable implementations.

However, the use of cyclic variations of common spreading code 38, whencombined with cyclic prefix 114 and processed as discussed below inreceiver 24, enables the mathematical communicative property.Consequently, despreading may now occur prior to equalization formultipath, thereby making effective equalization a relatively simpletask.

FIG. 13 shows a second embodiment of CDM to TDM converter 82 for usewith the fourth embodiment of the DSSS modulation section 44 (i.e., thetime domain embodiment). The digitized IF form of communication signal26, after being distorted through the communication channel, is appliedto a demultiplexer (DEMUX) 118 and a synchronization (SYNC) section 120.An output of synchronization section 120 feeds a cyclic prefix removalsection 122 of demultiplexer 118. Synchronization section 120 identifiesthe start of blocks 34, and cyclic prefix removal section 122 removesthe first-occurring p chips from each block 34. As discussed above, thelast-occurring p chips duplicate the first-occurring p chips, and thelast-occurring p chips and all other chips remain in each block 34. Thefirst-occurring p chips are removed because they are influenced bymultipath from the previous block 34 of communication signal 26. Allchips, which remain in each block 34, are influenced only by multipathfrom that block 34.

The block 34 of chips, with cyclic prefix 114 (FIG. 12) removed, passesto mismatched filter 100 for despreading and equalization. As discussedabove, due to the use of cyclic variations of spreading code 3 8 tospread symbols 3 6 and the inclusion of cyclic prefix 114, the matrixmultiplication which characterizes the channel now observes thecommunicative mathematical property. Consequently, despreading may occurbefore equalization.

Despreading may take place using a despreading code generator 124, adespreading section 126, and a combining section 128. Despreading codegenerator 124, despreading section 126, and combining section 128 areidentical in structure to spreading code generator 94, spreading section72, and combining section 78 in DSSS modulator section 44 of transmitter22 (FIG. 12), with the despreading code generated in despreading codegenerator 124 being related to spreading code 38. In particular,despreading code chips D_(n)=IFFT(1/FFT(C_(n))), where IFFT and FFTdenote inverse fast Fourier transform and fast Fourier transform,respectively, and C_(n) represents the chips of spreading code 38.

Spread substreams 130 are provided by demultiplexer 118 to multipliers132 in despreading section 126 along with the despreading code matrixfrom despreading code generator 124. The despreading code is applied inthe form of a cyclic Toeplitz matrix due to the use of cyclic variationsof a common spreading code to which the despreading code is related.Multipliers 132 provide despread substreams 134 to combiner 128 to adddespread substreams 134 on a chip by chip basis into a serialpre-composite baseband signal 136. Pre-composite baseband signal 136 isconverted into parallel symbol substreams 140 at a demultiplexer (DEMUX)13 8, and symbol substreams 140 are applied to a maximum likelihoodsequence estimation (MLSE) equalizer 142 or the equivalent. MLSEequalizer 142 may also be called a Viterbi equalizer. Parallel outputsfrom MLSE equalizer 142 feed a multiplexer (MUX) 144 which converts theparallel symbol substreams into baseband signal 84 for furtherprocessing by TDM demodulation section 86 (FIG. 3).

Those skilled in the art will appreciate that an MLSE equalizer is asimple structure, which is stable and can be effectively configured tocompensate for multipath. The coupling of MLSE equalizer 142 downstreamof despreading section 126 is possible due to the use of cyclicvariations of common spreading code 38 in transmitter 22 and cyclicprefix 114 to enable matrix multiplication exhibiting the mathematicalcommunicative property.

FIG. 14 shows a third embodiment of CDM to TDM converter 82 for use withthe fifth embodiment of DSSS modulation section 44 (i.e., the frequencydomain embodiment), discussed above in connection with FIG. 12. Thedigitized IF form of communication signal 26, after being distortedthrough the communication channel, is applied to demultiplexer (DEMUX)118 and synchronization (SYNC) section 120, as discussed above inconnection with FIG. 13. Likewise, cyclic prefix 114 (FIG. 12) isremoved at cyclic prefix removal section 122 of demultiplexer 118, asdiscussed above in connection with FIG. 13.

Spread substreams 13 0 are provided by demultiplexer 118 to atime-frequency domain transformation section 146, which complementstime-frequency domain transformation section 110 (FIG. 12). Thus, iftime-frequency domain transformation section 110 in DSSS modulationsection 44 implements an inverse fast Fourier transform (IFFT), thentime-frequency domain transformation section 146 desirably implements afast Fourier transform (FFT).

Mismatched filter 100 couples downstream of time-frequency domaintransformation section 146. In this third embodiment of CDM to TDMconverter 82 mismatched filter 100 may be implemented in a manner thatjoins despreading and equalization functions in a common frequencydomain equalizer. As illustrated in FIG. 14, coefficients for thefrequency domain equalizer take the form D*_(H(n))/D_(C(n)), whererepresents despreading code chips that are related to spreading code 38in the manner discussed above in connection with FIG. 13 and D*_(H(n))represents the complex conjugate of the transfer function of thechannel. One reason why a frequency domain equalizer is easy andeffective to implement is that coefficients are not proportional to theinverse of the transfer function of the channel. While despreading codechips are related to the inverse of the FFT of the spreading code, suchcoefficients do not pose problems because the designer controls the FFTof the code through code selection, and a spreading code having asubstantially flat spectral response may be selected, as discussed abovein connection with FIG. 9.

Parallel outputs of mismatched filter 100 pass in parallel to harddecision sections 148, and parallel outputs of hard decision sections148 are combined into serial baseband signal 84 in multiplexer (MUX) 144for further processing in TDM demodulation section 86 (FIG. 3).

Due to the enabling of the mathematical communicative property formatrix multiplication discussed above, mismatched filter 100 may residedownstream of time-frequency transformation section 146, which improvesthe efficacy and simplicity of the equalization and despreadingfunctions.

The present invention provides an improved method and apparatus forwireless communications. The present invention contemplates thecombination of TDMA and spread spectrum techniques so that wirelesscommunications capacity is increased over the capacities achievablethrough conventional TDMA and/or CDMA systems using an equivalent amountof spectrum. Furthermore, robust, simple, and inexpensive processingtechniques are usable in the present invention, making the presentinvention suitable for hubs, subscriber units, mobile stations, fixedstations, portable stations, and the like. The present invention may beadapted to and used in conjunction with a variety of modulation andmultiple access techniques, such as frequency division multiple access(FDMA) and orthogonal frequency division multiplexing (OFDM). Theadvantages and improvements of the present invention are achieved, atleast in part through the use of a composite RF communication signalwhich includes signal components obtained by modulating diverse branchesof a single user's data stream using cyclic variants of a commonspreading code. The present invention is configured to tolerateself-interference and is better able to tolerate multipath thanconventional DSSS-CDMA communication systems.

With reference to FIGS. 15 and 16, additional embodiments of the presentinvention will now be described. The embodiments described below employultra-wideband communication technology. Referring to FIGS. 15 and 16,ultra-wideband (UWB) communication technology employs discrete pulses ofelectromagnetic energy that are emitted at, for example, nanosecond orpicosecond intervals (generally tens of picoseconds to hundreds ofnanoseconds in duration). For this reason, ultra-wideband is oftencalled “impulse radio.” That is, the UWB pulses may be transmittedwithout modulation onto a sine wave, or a sinusoidal carrier, incontrast with conventional carrier wave communication technology. Thus,UWB generally requires neither an assigned frequency nor a poweramplifier.

Another example of sinusoidal carrier wave communication technology isillustrated in FIG. 15. IEEE 802.11a is a wireless local area network(LAN) protocol, which transmits a sinusoidal radio frequency signal at a5 GHz center frequency, with a radio frequency spread of about 5 MHz. Asdefined herein, a carrier wave is an electromagnetic wave of a specifiedfrequency and amplitude that is emitted by a radio transmitter in orderto carry information. The 802.11 protocol is an example of a carrierwave communication technology. The carrier wave comprises asubstantially continuous sinusoidal waveform having a specific narrowradio frequency (5 MHz) that has a duration that may range from secondsto minutes.

In contrast, an ultra-wideband (UWB) pulse may have a 2.0 GHz centerfrequency, with a frequency spread of approximately 4 GHz, as shown inFIG. 16, which illustrates two typical UWB pulses. FIG. 16 illustratesthat the shorter the UWB pulse in time, the broader the spread of itsfrequency spectrum. This is because bandwidth is inversely proportionalto the time duration of the pulse. A 600-picosecond UWB pulse can haveabout a 1.8 GHz center frequency, with a frequency spread ofapproximately 1.6 GHz and a 300-picosecond UWB pulse can have about a 3GHz center frequency, with a frequency spread of approximately 3.2 GHz.Thus, UWB pulses generally do not operate within a specific frequency,as shown in FIG. 15. In addition, either of the pulses shown in FIG. 16may be frequency shifted, for example, by using heterodyning, to haveessentially the same bandwidth but centered at any desired frequency.And because UWB pulses are spread across an extremely wide frequencyrange, UWB communication systems allow communications at very high datarates, such as 100 megabits per second or greater.

Also, because the UWB pulses are spread across an extremely widefrequency range, the power sampled in, for example, a one megahertzbandwidth, is very low. For example, UWB pulses of one nano-secondduration and one milliwatt average power (0 dBm) spreads the power overthe entire one gigahertz frequency band occupied by the pulse. Theresulting power density is thus 1 milliwatt divided by the 1,000 MHzpulse bandwidth, or 0.001 milliwatt per megahertz (−30 dBm/MHz).

Generally, in the case of wireless communications, a multiplicity of UWBpulses may be transmitted at relatively low power density (milliwattsper megahertz). However, an alternative UWB communication system maytransmit at a higher power density. For example, UWB pulses may betransmitted between 30 dBm to −50 dBm.

Several different methods of ultra-wideband (UWB) communications havebeen proposed. For wireless UWB communications in the United States, allof these methods must meet the constraints recently established by theFederal Communications Commission (FCC) in their Report and Order issuedApr. 22, 2002 (ET Docket 98-153). Currently, the FCC is allowing limitedUWB communications, but as UWB systems are deployed, and additionalexperience with this new technology is gained, the FCC may expand theuse of UWB communication technology. It will be appreciated that thepresent invention may be applied to current forms of UWB communications,as well as to future variations and/or varieties of UWB communicationtechnology.

For example, the April 22 Report and Order requires that UWB pulses, orsignals occupy greater than 20% fractional bandwidth or 500 megahertz,whichever is smaller. Fractional bandwidth is defined as 2 times thedifference between the high and low 10 dB cutoff frequencies divided bythe sum of the high and low 10 dB cutoff frequencies. However, theserequirements for wireless UWB communications in the United States maychange in the future.

Communication standards committees associated with the InternationalInstitute of Electrical and Electronics Engineers (IEEE) are consideringa number of ultra-wideband (UWB) wireless communication methods thatmeet the current constraints established by the FCC. One UWBcommunication method may transmit UWB pulses that occupy 500 MHz bandswithin the 7.5 GHz FCC allocation (from 3.1 GHz to 10.6 GHz). In oneembodiment of this communication method, UWB pulses have about a2-nanosecond duration, which corresponds to about a 500 MHz bandwidth.The center frequency of the UWB pulses can be varied to place themwherever desired within the 7.5 GHz allocation. In another embodiment ofthis communication method, an Inverse Fast Fourier Transform (IFFT) isperformed on parallel data to produce 122 carriers, each approximately4.125 MHz wide. In this embodiment, also known as Orthogonal FrequencyDivision Multiplexing (OFDM), the resultant UWB pulse, or signal isapproximately 506 MHz wide, and has a 242 nanosecond duration. It meetsthe FCC rules for UWB communications because it is an aggregation ofmany relatively narrow band carriers rather than because of the durationof each pulse.

Another UWB communication method being evaluated by the IEEE standardscommittees comprises transmitting discrete UWB pulses that occupygreater than 500 MHz of frequency spectrum. For example, in oneembodiment of this communication method, UWB pulse durations may varyfrom 2 nanoseconds, which occupies about 500 MHz, to about 133picoseconds, which occupies about 7.5 GHz of bandwidth. That is, asingle UWB pulse may occupy substantially all of the entire allocationfor communications (from 3.1 GHz to 10.6 GHz).

Yet another UWB communication method being evaluated by the IEEEstandards committees comprises transmitting a sequence of pulses thatmay be approximately 0.7 nanoseconds or less in duration, and at achipping rate of approximately 1.4 giga pulses per second. The pulsesare modulated using a Direct-Sequence modulation technique, and iscalled DS-UWB. Operation in two bands is contemplated, with one band iscentered near 4 GHz with a 1.4 GHz wide signal, while the second band iscentered near 8 GHz, with a 2.8 GHz wide UWB signal. Operation may occurat either or both of the UWB bands. Data rates between about 28Megabits/second to as much as 1,320 Megabits/second are contemplated.

Thus, described above are three different methods of wirelessultra-wideband (UWB) communication. It will be appreciated that thepresent invention may be employed using any one of the above-describedmethods, variants of the above methods, or other UWB communicationmethods yet to be developed.

Certain features of the present invention may be employed by anultra-wideband (UWB) communication system. For example, one embodimentof an UWB communication system divides a stream of data conveyingsymbols into a plurality of unspread substreams. A common spreading codeis generated at the ultra-wideband transmitter, and each of the unspreadsubstreams are spread using the common spreading code to form aplurality of spread substreams. The spread substreams are combined toform a composite signal that is transmitted using a plurality ofdiscrete electromagnetic pulses.

In another embodiment, an ultra-wideband communication system comprisesa demultiplexer for dividing a stream of data conveying symbols into aplurality of unspread substreams. A spreading section is coupled to thedemultiplexer and configured to generate a plurality of spreadsubstreams from the plurality of unspread substreams. A combiningsection is coupled to the spreading section and configured to form acomposite signal from the plurality of spread substreams, and atransmission section is coupled to the combining section and configuredto transmit the composite signal over an ultra-wideband communicationchannel.

The UWB devices, systems and/or methods in the above-describedembodiments communicate with each other by transmitting and receiving aplurality of discrete electromagnetic pulses, as opposed to asubstantially continuous carrier wave. Each pulse may have a durationthat can range between about 10 picoseconds to about 1 microsecond, anda power that may range between about +30 dBm to about −60 dBm, asmeasured at a single frequency.

The present invention may be employed in any type of network, be itwireless, wire, or a mix of wire media and wireless components. That is,a network may use both wire media, such as coaxial cable, and wirelessdevices, such as satellites, or cellular antennas. As defined herein, anetwork is a group of points or nodes connected by communication paths.The communication paths may use wires or they may be wireless. A networkas defined herein can interconnect with other networks and containsub-networks. A network as defined herein can be characterized in termsof a spatial distance, for example, such as a local area network (LAN),a personal area network (PAN), a metropolitan area network (MAN), a widearea network (WAN), and a wireless personal area network (WPAN), amongothers. A network as defined herein can also be characterized by thetype of data transmission technology used by the network, such as, forexample, a Transmission Control Protocol/Internet Protocol (TCP/IP)network, a Systems Network Architecture network, among others. A networkas defined herein can also be characterized by whether it carries voice,data, or both kinds of signals. A network as defined herein may also becharacterized by users of the network, such as, for example, users of apublic switched telephone network (PSTN) or other type of publicnetwork, and private networks (such as within a single room or home),among others. A network as defined herein can also be characterized bythe usual nature of its connections, for example, a dial-up network, aswitched network, a dedicated network, and a non-switched network, amongothers. A network as defined herein can also be characterized by thetypes of physical links that it employs, for example, optical fiber,coaxial cable, a mix of both, unshielded twisted pair, and shieldedtwisted pair, among others.

The present invention may be employed in any type of wireless network,such as a wireless PAN, LAN, MAN, or WAN. In addition, the presentinvention may be employed in wire media, as the present inventiondramatically increases the bandwidth of conventional networks thatemploy wire media, such as hybrid fiber-coax cable networks, or CATVnetworks, yet it can be inexpensively deployed without extensivemodification to the existing wire media network.

Thus, it is seen that systems and methods of ultra-widebandcommunications are provided. One skilled in the art will appreciate thatthe present invention can be practiced by other than the above-describedembodiments, which are presented in this description for purposes ofillustration and not of limitation. For example, those skilled in theart will appreciate that the order of time-frequency domaintransformation and spreading functions may be reversed from that shownin FIG. 12. The specification and drawings are not intended to limit theexclusionary scope of this patent document. It is noted that variousequivalents for the particular embodiments discussed in this descriptionmay practice the invention as well. That is, while the present inventionhas been described in conjunction with specific embodiments, it isevident that many alternatives, modifications, permutations andvariations will become apparent to those of ordinary skill in the art inlight of the foregoing description. Accordingly, it is intended that thepresent invention embrace all such alternatives, modifications andvariations as fall within the scope of the appended claims. The factthat a product, process or method exhibits differences from one or moreof the above-described exemplary embodiments does not mean that theproduct or process is outside the scope (literal scope and/or otherlegally-recognized scope) of the following claims.

1. A method of ultra-wideband communication, comprising: dividing astream of data conveying symbols into a plurality of unspread substreamsat an ultra-wideband transmitter; generating a common spreading code atthe ultra-wideband transmitter; spreading each of the unspreadsubstreams using the common spreading code to form a plurality of spreadsubstreams; combining the plurality of spread substreams to form acomposite signal; and transmitting the composite signal from theultra-wideband transmitter.
 2. The method of claim 1, wherein the stepof transmitting the composite signal from the ultra-wideband transmittercomprises transmitting a plurality of discrete electromagnetic pulses,with each pulse having a duration that can range between about 10picoseconds to about 1 microsecond.
 3. The method of claim 1 furthercomprising: receiving the composite signal at an ultra-widebandreceiver; and despreading the composite signal at the ultra-widebandreceiver using a mismatched filter to generate a baseband signal; wherethe mismatched filter comprises a matched filter and a sidelobesuppression filter.
 4. The method of claim 1, wherein the spreadingactivity comprises temporally offsetting the spreading of each of theunspread substreams so that the spread substreams correspond to unspreadsubstreams spread by cyclical variations of the common spreading code.5. The method of claim 1, wherein the dividing of the stream of dataconveying symbols produces successive blocks of symbols, where eachblock of symbols is comprised of symbols concurrently present in eachunspread substream, each block having a block period, and the temporallyoffsetting and the combining of substreams are configured so that for aportion of each block period the composite signal is responsive tosymbols from more than one block.
 6. The method of claim 1, wherein thedividing of the stream of data conveying symbols produces successiveblocks of symbols, where each block of symbols is comprised of symbolsconcurrently present in each unspread substream, each block having ablock period, and the temporally offsetting and the combining ofsubstreams are configured so that for a portion of each block period thecomposite signal is responsive to symbols from only common blocks. 7.The method of claim 3, wherein the step of dividing of the stream ofdata conveying symbols and the spreading each of the unspread substreamsproduce M unspread substreams and M spread substreams, where M is aninteger number; and wherein the step of combining combines the M spreadsubstreams so that, for each block P of the M spread substreams occurfirst in the composite signal, where P is an integer number less than M;and wherein the step of combining additionally comprises, for eachblock, repeating the P spread substreams so that the P spread substreamsalso occur last in the composite signal; and where between the step ofreceiving and despreading the composite signal, a step of removing thefirst-occurring P spread substreams from the communication signal isperformed.
 8. The method of claim 1, further comprising applying blockencoding to an input stream of data so that encoding blocks coincidewith said successive blocks of symbols.
 9. The method of claim 3,further comprising equalizing the composite signal at the ultra-widebandreceiver following the despreading activity using a maximum likelihoodsequence estimation (MLSE) equalizer.
 10. The method of claim 3, furthercomprising: performing a first time-frequency domain transformation onthe unspread substreams at the ultra-wideband transmitter prior to thespreading activity; and performing a second time-frequency domaintransformation on the composite signal at the ultra-wideband receiverprior to the despreading activity.
 11. The method of claim 10, furthercomprising equalizing the communication signal at the ultra-widebandreceiver following the second time-frequency domain transformation usinga frequency domain equalizer.
 12. The method of claim 4, wherein thecyclical variations of the common spreading code are applied to theunspread substreams as a matrix in a cyclic Toeplitz form.
 13. Themethod of claim 3, wherein the ultra-wideband receiver is a firstultra-wideband receiver, the mismatched filter is a first mismatchedfilter, the baseband signal is a first baseband signal, and the methodadditionally comprises: receiving the composite signal at a secondultra-wideband receiver; despreading the composite signal in the secondultra-wideband receiver using a second mismatched filter to generate asecond baseband signal; generating the stream of data-conveying symbolsas a time division multiple access (TDMA) stream having a plurality oftime slots wherein a first one of the plurality of time slots isassigned to the first ultra-wideband receiver and a second one of theplurality of time slots is assigned to the second ultra-widebandreceiver; evaluating the first baseband signal at the firstultra-wideband receiver to detect the first one of the time slots; andevaluating the second baseband signal at the second ultra-widebandreceiver to detect the second one of the time slots.
 14. The method ofclaim 1, wherein the ultra-wideband transmitter is a firstultra-wideband transmitter and the composite signal is a first compositesignal, the first transmitter is configured so that the first compositesignal substantially occupies a predetermined spectrum and the firstcomposite signal is detectable throughout a first radio coverage area,further comprising the step of: transmitting a second composite signalfrom a second ultra-wideband transmitter, the second composite signalsubstantially occupying the predetermined spectrum and being detectablethroughout a second radio coverage area which is adjacent to the firstradio coverage area.
 15. An ultra-wideband communication systemcomprising: a demultiplexer for dividing a stream of data conveyingsymbols into a plurality of unspread substreams; a spreading sectioncoupled to the demultiplexer and configures to generate a plurality ofspread substreams from the plurality of unspread substreams; a combiningsection coupled to the spreading section and configured to form acomposite signal from the plurality of spread substreams; and atransmission section coupled to the combining section and configured totransmit the composite signal over an ultra-wideband communicationchannel.
 16. The system of claim 15, wherein the composite signaltransmitted by the transmission section comprises a plurality ofdiscrete electromagnetic pulses, with each pulse having a duration thatcan range between about 10 picoseconds to about 1 microsecond.
 17. Thesystem of claim 15, wherein each of the plurality of spread substreamscorresponds to each of the plurality of unspread substreams modulated bya common spreading code.
 18. The system of claim 15, wherein each of theplurality of spread substreams corresponds to each of the plurality ofunspread substreams modulated by cyclic variations of a common spreadingcode.
 19. The system of claim 15, wherein the composite signal is afirst composite signal, with the ultra-wideband communication systemfurther comprising: a receiving section configured to receive a secondcomposite signal from the ultra-wideband communications channel; and adespreading section coupled to the receiving section, the despreadingsection being configured to generate a baseband signal in response tothe second composite signal.
 20. The system of claim 15, wherein thedemultiplexer produces successive blocks of symbols, where each blockincludes symbols concurrently present in each unspread substream, andeach block has a block period, and wherein the spreading section isconfigured so that for a portion of each block period the compositesignal is responsive to symbols from two different blocks.
 21. Thesystem of claim 15, further comprising a time division multiple access(TDMA) modulation section coupled to the demultiplexer, the TDMAmodulation section being configured so that the composite signal is aTDMA signal for which recipients are distinguished from one another bybeing assigned to different time slots.
 22. The system of claim 15,wherein the transmission section is a first transmission section, andthe composite signal is a first composite signal, and the ultra-widebandcommunication system has a first and a second adjacent radio coverageareas, and the ultra-wideband communication system further comprises: asecond transmission section configured to transmit a second compositesignal to the second adjacent radio coverage area over an ultra-widebandcommunication channel, the first and the second composite signals beingtransmitted using a common spectrum.
 23. The system of claim 19,wherein: the demultiplexer produces successive blocks of M symbols thatconcurrently influence the unspread substreams, where M is an integernumber; the spreading section is configured to produce M spreadsubstreams, the M spread substreams being responsive to symbols onlyfrom common blocks; the combining section is configured so that, foreach block, P of the M spread substreams occur first in the compositesignal, where P is an integer number less than M; the combining sectionis further configured so that, for each block, the P spread substreamsare repeated so that the P spread substreams also occur last in thecomposite signal; and wherein the ultra-wideband communication systemfurther comprises: a cyclic prefix removal section coupled between thereceiving and the despreading sections for removing the first-occurringP spread substreams from the composite signal.
 24. The system of claim19, further comprising a maximum likelihood sequence estimation (MLSE)equalizer communicating with the despreading section.
 25. The system ofclaim 19, further comprising: a first time-frequency domaintransformation section coupled between the demultiplexer and thespreading section; a second time-frequency domain transformation sectioncoupled between the receiver and the despreading section, wherein thedespreading section comprises a frequency domain equalizer.